The best mixers for many applications are diode ring mixers. Transistors and field-effect transistors may also be used as semiconductor switches in mixer designs, but require more components to drive them, and introduce more design complications.
Referring to FIGS. 1a, 1b, 1c, 1d and 1e the process of modulation, as known in the art, is described. FIG. 1a identifies a continuous signal, f.sub.O, on amplitude and time axes, and FIG. 1b represents a carrier signal, f.sub.LO, on the same axes. Mixing these two signals together, also known as modulating or heterodyning, results in the modulated signal shown as f.sub.MOD in FIG. 1d. The modulated signal, f.sub.MOD, has the frequency of the carrier signal, f.sub.LO, and the amplitude of the continuous signal, f.sub.O.
Although this example will be explained with respect to an input signal, f.sub.O, composed of a single frequency, it is understood that the process applies in the same manner to signals with either multiple frequencies, or complete spectra of frequencies. The theory of upper and lower sidebands, for example, is known in the art and will not be reviewed herein.
Diode mixers are one of a group of semiconductor switching mixers called "chopper" mixers. These mixers couple or "chop" segments of the continuous input signal to the output in time with the carrier frequency, and do not allow the remainder of the continuous input signal to pass to the output. A semiconductor switch which is biased to couple the continuous signal to the output when the carrier frequency f.sub.LO has a positive magnitude would provide an output as shown in FIG. 1c. During each positive half-cycle of the carrier frequency, f.sub.LO, the semiconductor switch will allow the continuous signal to pass to the output, giving an output signal S(t).
As an electromagnetic signal is composed of sinusoidal waves, these segments will be composed of sinusoids at a fundamental frequency and smaller amplitude sinusoids at harmonic frequencies of the fundamental. In addition to the fundamental and harmonic signal components, the non-linearities of chopper mixers in the prior art also produce intermodulation products. These intermodulation products consume signal power and may interfere with the target signals.
The process described thus far is for a single mixer. A double balanced mixer has a second semiconductor switch, which chops and inverts complementary segments of the continuous input signal, as shown in FIG. 1e. These inverted segments have the same sinusoid pattern as the output of the single mixer, resulting in a more powerful output signal.
Modulation is used in basically the same manner as described above in a broad range of applications: televisions, microwave communications and spectrum analysers, as well as modulating radio frequency (RF) signals into intermediate frequency (IF) signals in radio receivers.
A circuit for performing such double mixing, as known in the art, is shown in FIG. 2. This diode ring mixer circuit 10 has an input for a continuous signal 12, an input for a local oscillator signal 14, an output for a modulated signal 16, a signal transformer 18, a local oscillator transformer 20, and four diodes 22, 24, 26, and 28.
The local oscillator input 14, local oscillator transformer 20 and diodes 22, 24, 26, and 28 alternately couple one end of the primary winding of the signal transformer 18 to ground, and then the other. This allows the input 12 to alternately conduct through upper and lower halves of the primary winding of the signal transformer 18. This gives a chopping and an inverted chopping of the input signal 12 as described with respect to FIGS. 1a, 1b, 1c, 1d and 1e above.
The local oscillator components cause this chopping by alternately forward-biasing diode pair 22 and 24, and diode pair 26 and 28. Or put another way, the local oscillator components cause the diodes to act as switches. During a positive half cycle of the local oscillator at input 14, a corresponding positive half cycle would pass through the local oscillator transformer 20, and create a positive potential difference across points A and C of the mixer 10. This positive potential difference would cause diodes 26 and 28 to be forward-biased, thus conducting, and diodes 22 and 24 to be reverse-biased, thus non-conducting.
With diodes 26 and 28 conducting, two circuits are completed. Because diodes 26 and 28 are conducting and balanced to the grounded center tap of the local oscillator transformer 20, point B is essentially at ground potential. Firstly, the circuit of the local oscillator signal from both halves of the secondary of the local oscillator transformer 20 is completed to the ground potential at point B. Since this circuit is complete, the local oscillator signal does not pass through to the signal transformer 18. Secondly, with point B at ground potential, the input signal from input 12 can flow through the lower half of the signal transformer 18 to ground, coupling the continuous signal input 12 to the output 16, and allowing a segment of the continuous input signal to pass through the signal transformer 18 to the output 16 without phase reversal. With both diodes 26 and 28 in forward-bias, the continuous signal current from input 12, flows to ground via the two halves of the secondary winding of the local oscillator transformer 20 in opposite directions, so there is no net magnetization of the local oscillator transformer 20 core, and the transformer 20 offers no impedance to the signal.
Similarly, during a negative half cycle of the local oscillator at input 14, a corresponding negative half cycle would pass through the local oscillator transformer 20, and cause a negative potential difference across points A and C of the mixer 10. This negative potential difference would cause diodes 22 and 24 to be in forward-bias, thus conducting, and diodes 26 and 28 to be in reverse-bias, thus non-conducting.
With diodes 22 and 24 conducting, again two circuits are completed. Firstly, the local oscillator signal from the secondary of the local oscillator transformer 20 is completed, and the local oscillator signal does not pass through to the signal transformer 18. Secondly, as described above, with diodes 22 and 24 now behaving as closed circuits, point D is essentially at ground potential. With point D at ground potential, the input signal 12 can flow through the upper half of the signal transformer 18 to ground, causing an inverted segment of the input signal from input 12 to pass through the signal transformer 18 to the output 16.
In this way, the input signal is modulated as described with respect to FIGS. 1a, 1b, 1c, 1d and 1e. The problem with this circuit is that when either pair of diodes is forward-biased, the voltage drop across A and C will be limited to the voltage drop across the forward-biased diode pair plus the amplitude of the input signal 12, or V.sub.s. Therefore, the reverse-bias voltage applied to the non-conducting diode pair will be limited to that drop across the forward-biased diode pair plus V.sub.s. With typical Schottky signal diodes, this is about 400 mV.+-.V.sub.s. Because 400 mV is not a great deal larger than the input signal V.sub.s, even small variances in the input signal V.sub.s will cause intermodulation products to be generated and output at 16. It is believed that the dominant mechanisms are that junction capacitance and reverse leakage of the diodes are non-linear functions of voltage.
If these intermodulation products fall inband, they may degrade the output signal itself, or if they fall out of band, they may appear as spectral spreading or interfere with other signals. These effects are typically important considerations in the design of transmitters and receivers for use in wireless systems or CATV modems.
In Frequency Division Multiple Access systems (FDMA), for example, a receiver may wish to listen to a distant transmitter in one channel, but be overwhelmed by intermodulation products produced by a nearby transmitter radiating in an adjacent channel. This is known as the "near-far" problem.
In Code Division Multiple Access systems (CDMA), methods are known to reduce the effect of noise created by intermodulation products, but such methods require overheads of transmitted code sequences or computational analysis of timing, carrier phase, or other parameters. The greater the noise level, the greater the overhead necessary to compensate. Reducing the noise level allows for reduced compensatory overheads and higher efficiency.
In receivers, front end components such as preselection filters, attenuators, and front-end Automatic Gain Control (AGC) systems are needed to protect mixers from excessive signal levels. Receivers may also pick up noise in out of band frequencies and inadvertently create intermodulation products of that noise which fall inband. In transmitters, the ability to operate at higher signal levels can permit a wider choice of architectures.
Emerging demands to provide data transmission over existing coaxial cable infrastructures currently carrying television signals, and existing twisted pair infrastructures currently carrying voice telephone services, require digital modems with greater speed and reduced noise levels. An improved mixer design would allow digital modems to operate at higher speeds with less overhead to compensate for noise levels.
There is therefore a need for a double balanced diode mixer which provides reduced levels of intermodulation products. This design must be provided with consideration for the cost of electrical components, circuit manufacturing and physical board area.